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Ideas for Design: Quad Video Amp Splits And Buffers An S-Video Signal | 4-To-20-mA Loop Powers Temperature Sensor

Series-Connected Transistors Use Differential Heating To Sense Airflow

by Stephen Woodward
University of North Carolina, Venable Hall, CB3300, Chapel Hill, NC 27599-3300
e-mail: woodward@.unc.edu

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Z_T = Z_J + 1/(S_C + K_T √(A_F))

where: Z_J = "total immersion" junction-to-case thermal impedance = 44°C/W

S_C = still-air case-to-ambient conductivity = 6.4 mW/°C

K_T = King's Law thermal diffusion constant = 75 µW/°C-√(fpm)

A_F = airspeed in ft/min

In this model, the raw sensor output is inherently nonlinear with airspeed, a problem common to all thermal airspeed sensors. To compensate, thermal anemometer designs must include some provision for measurement linearization. The circuit in Figure 1 combines ideas from two earlier IFDs ("Low-Power Thermal Airspeed Sensor," electronic design, May 25, 1998, p. 116; and "Low-Power Solid-State Airflow Detector," electronic design, Jan. 22, 2001, p. 118). In doing so, it implements a simple, linearized (±5%), ambient-temperature-compensated thermal anemometer. A robust, power-efficient device, it draws less than 1 W of total operating power from a single regulated 5-V rail.

In operation, A1 maintains a constant temperature differential (about 25°C) between Q1 and Q2, independent of changes in thermal impedance and ambient temperature. A1 achieves this by maintaining a constant ratio between the two transistors' V_BE voltages. It can do so by controlling the collector currents of the series-connected devices, and thereby their power dissipation.

Since both transistors pass the same current (I_Q), their relative power dissipations are determined solely by their respective V_CE voltages. V_Q1 > V_Q2 for all valid operating modes (Q3 sees to this). So for any given I_Q, Q1 will always dissipate more power and more heat, meaning it will run hotter than diode-connected Q2. Consequently, as airflow increases and thermal impedance decreases, A1 can hold any chosen Q1/Q2 temperature differential by increasing I_Q. The resulting airflow-dependent I_Q is sensed by R1, then offset and boosted by A2. In turn, it becomes the 0- to 2.5-V anemometer output signal V_O, scaled for 10 mV = 1 fpm = 0 to 250 fpm (~2.5 kts).

Meanwhile, Q3 acts with A2 to limit the maximum voltage across R1 to about 2 V. This is done to avoid the risk of latch-up, which would occur if A1's output were allowed to rise too near the 5-V rail. In that event, V_Q1 would approach V_Q2. As a result, it would be impossible to achieve the programmed temperature differential and Q1/Q2 V_BE ratio, no matter how high I_Q might rise. Similarly, R7 and R8 prevent latch-up when the circuit is first powered up.

But what about measurement linearization? As illustrated in Figure 2, the inherent quadratic relationship that exists between I_Q and Q1/Q2's power dissipation does a fair job of canceling nonlinearity.

2. The quadratic relationship between I_Q and Q1/Q2's power dissipation does a fair job of canceling nonlinearity, erasing all but ±5% FSR linearity error over the range of on-scale airspeeds.

It erases all but ±5% FSR linearity error over the entire range of on-scale airspeeds. Also, anemometer calibration is quick and straightforward. The transistor sensor pair is simply placed in slowly moving air (almost, but not quite stagnant; A_F = 5 to 7 fpm is ideal). R6 is then adjusted for V_O = 0.

The "tranemometer" is illustrated with circuit constants that scale its output for V_O = 0.01 V/fpm = 1 V/kt. Yet virtually any range of airflow rates can be accommodated with appropriate choices for R1, R2, and R3.

Ideas for Design: Quad Video Amp Splits And Buffers An S-Video Signal | 4-To-20-mA Loop Powers Temperature Sensor

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