Rick Campbell's R2 receiver prompted the question, "Can it be prodded to perform even better?". A good deal of simulation, and breadboarding showed a few minor flaws, some of which Rick has already corrected. My main interest is in improving the audio chain. The goal was to achieve an overall noise figure 10dB or under. A new method of tuning the audio phase shifter is introduced that allows sideband suppression of 50 dB. A lower noise audio preamp, a sharp, tunable audio filter, and a more powerful no-adjustment audio driver are included as well.
Low-level Audio
Preamp
The old common-base audio preamp, having
a noise figure of 5dB is an obvious target to improve the noise figure.
A common-emitter configuration, with shunt feedback gives 50 ohms input
resistance (preferred by the mixer) and a 2dB noise figure. Two paralleled
2N4401 transistors results in better noise performance at such a low input
impedance. A gain of 60 masks most of the noise in the op-amp phase shifter
that follows. Common collector ouput stages provide a low output impedance
needed to drive the audio phase shifter.
A simpler diplexer between mixer and
preamp shaves a little off the noise figure too. Rick's R2 diplexer has
steeper stopband slope, but about 2dB loss. The TOKO 10RB inductors
simply have too much series resistance: I wound my own on small potcores,
14x8 mm. The potcore bobbins are very easy to wind, but its tricky to get
the inductance exact - too much pressure from a mounting screw can change
inductance dramatically. Potcore mounting hardware from the manufacturer
is recommended. The 1.2mH inductor should have low internal capacitance
- 35 turns #32 wire in a single layer on a FT37-77 ferrite toroid.
Diplexer low-pass frequency
at 6700 Hz., and high-pass frequency at 170 Hz are outside the audio passband
(350 - 3500 Hz.). This means that unwanted phase shifts from mismatched
diplexer parts are less troublesome, relaxing the need for extremely accurate
component values. With the tuning method described below, some of the bad
effects of mismatched components can be tuned out. Nevertheless, diplexer
components were matched with the aid of a commercial Maxwell impedance
bridge.
Quadrature Phase
Shifter
The phasing R's and C's were all scaled
so that resistor values were much smaller. Otherwise, the Johnson noise
from those warm, large value resistors adds so much noise that preamp gain
would have to be much higher to achieve the same noise figure. A return
to a low- noise bipolar op-amp (LM837) gives lower noise than the FET-input
TL074 with these lower resistor values. The Motorola MC33079P should work
equally well.
Most low-noise op-amps have
extended frequency response as a side-effect. My circuit layout was slightly
unstable, oscillating at 3 Mhz. Running at low power supply voltage doesn't
help, but oscillations are most likely caused by too much stray capacitance
between the inputs and ground. A 10pF feedback capacitor is usually enough
to correct the problem, and doesn't add noticeable phase error. A simpler
USB/LSB sideband switching arrangement means a few less op-amps.
This phasing circuit is capable of high
sideband suppression; a minimum of 58 dB (figure 1). As Rick has mentioned,
actually doing this well is very very difficult, mostly because of component
tolerances. Capacitors C24, C25, C26, C27, C28, C30 should all be
of the same type. The Panasonic P-Series (polypropylene) capacitors have
tight tolerance, and good temperature stability. Philips 460 series are
even better, but unavailable in the larger sizes. Polystyrene capacitors
are good too, but only small values are available. Almost all the resistors
should be 1% precision resistors.
Trimming these components
is considered by many to be too difficult, partly because of the need for
a quadrature signal generator. I'm a believer in building
your own test equipment; a very simple and accurate quadrature generator
was developed that uses two common CMOS chips. This circuit can be built
temporarily on a protoboard to trim the three phasing trimpots R43, R47,
and R51 for best alternate sideband suppression.
The signal generator must
supply two signals of the same frequency, same amplitude, but with exactly
a 90 degree phase relationship. Since each of the three trimpots to be
tuned affects some audio frequencies more than others, these two signals
must be frequency agile: 300 - 4000 Hz. Two square waves in quadrature
are easily generated with a logic shift register. However, the harmonics
of those square waves make it very difficult to determine the null
point for the fundamental frequency of interest. Sharp low-pass filtering
in the following audio stages can eliminate those harmonics. Simply listen
for minimum signal of the resulting sine wave.
U7 is a simple square-wave
oscillator whose frequency can be adjusted with R76. It oscillates at four
times the output frequency. The dual flip-flop U6 is connected as a two-bit
shift register. Outputs at pin 1 and 13 are at the same frequency, but
shifted 90 degrees in phase. R72 - R74 divide the output voltage down to
a small value (with a 50 ohm output impedance) that the preamps can handle.
Since there is so much gain in the R2's audio chain, the possibility of
ground loops can adversely affect trimming adjustments. Power the signal
generator from a separate power supply, perhaps a 9v battery. It draws
less than a few milliamps.
You might wish to set the
three trimpots with an accurate ohmmeter first. Theoretical values of resistances
from the non-inverting inputs of op-amp U5b, U5a, U4b to 6V COMMON are
listed on the schematic diagram in curly braces.
Apply the two outputs of
the quadrature generator to the two diplexer inputs. Now adjust the SCAF
frequency (R15) to pass only the fundamental frequency, and eliminate all
the harmonics. You can do this by ear, or look at the voltage at C5 with
an oscilloscope. You should be able to adjust R15 to hear (or see) a sine
wave. Flip the USB/LSB switch to the side giving the lowest amplitude.
Adjust R29 for minimum signal. Now you can go to the three 500 ohm phasing
trimpots.
Iterative tuning will be
required, since each trimpot affects the setting of its neighbor a little.
R51 will trim the highest audio frequencies: try nulling for best sideband
rejection at 3.4 KHz. R47 trims mid-frequencies: try for a null at 715
Hz. R43 trims the lower end: 295 Hz. At each of these frequencies, adjust
the SCAF cutoff frequency to pass most of the fundamental frequency, but
reject harmonics of the square wave. Rock R29 back and forth as well to
help find the best null. Make sure that you get deep nulls at all six frequencies
in figure 1.They are at 295 Hz, 417 Hz, 715 Hz, 1340
Hz, 2370 Hz, and 3380 Hz.
Mismatching in the local
oscillator quadrature hybrid, and an unbalanced R.F. splitter can easily
degrade sideband rejection shown in figure 1.
The butterworth band-pass
filter following the audio phasing combiner U4d passes audio from 350 Hz
to 3500 Hz. Again, the TOKO 120 mH pre-wound coils have been replaced
with hand-wound potcores. The 120 mH 10RB TOKO coils have a Q of three
at low audio frequencies...a little too low to be useful. However the R10B
type 33mH coils have acceptable Q for the low-pass part of this filter.
Bandpass filter components need not have high accuracy.
Audio Amplifier
From the volume control, Q6 amplifies
the audio signal by about 100. Good noise performance is needed here, especially
at low volume. The 2N4403 PNP transistor gives a noise figure of about
1dB, with a 500 ohm source resistance, and biased at about 1ma. Only exotic,
expensive low-noise op-amps can give better performance than this $0.16
transistor. However, the integrated circuit amplifiers are often
better at rejecting noise and hum present on the power supply.Q5 is added
to isolate Q6 from power supply hum, noise and feedback.
A non-critical op-amp follows
(U1a) with a gain of 16, and provides nearly rail-to-rail output voltage
swing.
Switched Capacitor Audio Filter (S.C.A.F)
U2
These devices are so easy to apply -
they're impossible to pass up. And the variable lowpass tuning that they
make possible is a great bonus. Unfortunately, these devices are relatively
noisy, necessitating their use at high amplitudes. You have a choice
of pin compatible filters to plug in here:
MAX292 - Bessel filter for best rise & fall shape with no
ringing whatsoever. (not recommended)
MAX291 - Butterworth filter for flat frequency response and
a little ringing. (recommended)
MAX293 - Elliptic filter for steep stopband slope but more ringing.
(not recommended)
MAX294 - Elliptic filter giving even steeper slope (not recommended)
All are available from Digi-Key for about $6.
Instead of using the built-in oscillator, the SCAF filter chip is driven from an external variable-frequency oscillator (R15, a front-panel variable resistor sets the frequency). U3 oscillates at exactly 100 times the cutoff frequency, providing a continuously variable filter from 350 Hz to 4000 Hz. This oscillator can be gated on and off with one of the CMOS logic inputs. With no clock, the SCAF filter stops in its tracks, holding its output voltage constant. This makes an extremely clean mute. R16, C8 and D3 have a fast attack, and slow decay (about 10msec) appropriate for break-in keying. The R16-C8 time constant can be easily changed if you need a mute with a longer tail.
Power Amplifier
Because the SCAF filter works best with
high-level signals, the power amplifier needs little voltage gain, but
lots of current gain. All the common integrated circuit power amps have
too much voltage gain to be useful here. Q3, Q4, Q1, Q2 have a composite
voltage gain of one.This circuit self-biases to a quiescent current (class
AB) of about 16 ma.
Since the SCAF filter must run from
a lower supply voltage, a tiny bit of gain is needed. Op- amp U1b supplies
this gain, and reduces distortion to a very low level. The combination
of a rail-to-rail op-amp with the bootstrapped output stage (C2, C3) result
in an output swing nearly equal to the supply voltage. Be warned: with
no AGC this amplifier will make you jump when an unexpected QRO signal
arrives in the passband.
The high currents drawn by the amplifier
must be routed carefully to avoid howling oscillations. Ground loops are
difficult to avoid when dealing with such high overall audio gains. The
collectors of Q1, Q2, Q3, Q4 and resistors R3, R6 should be connected to
the supply voltage separately from the rest of the receiver. The "ground"
lead of the speaker should be connected to exactly the same point as the
collector of Q2 and Q4. A grounded phono plug connection to the speaker
is asking for trouble, if the chassis is attached to any other part of
the receiver. Fortunately, the SBL-1 mixer ground-isolates the local oscillator
and radio-frequency input stages, so you shouldn't have to worry about
ground-isolating these inputs. A headphone jack, if used, should be ground-isolated
too.
The LMC662 CMOS op-amp sets
an upper power supply limit of 16v. High performance eclipsed low-power
operation as a design goal - 100mA total current is drawn from the supply
- more at high audio levels.
Conclusions
High-gain audio amplification
needed by direct conversion receivers will always be difficult to deal
with. Microphonics in the low-level preamp are a problem. Hum pickup by
the diplexer's magnetic components and ground loops are problems too: a
ferrite-free diplexer is a tempting future project. While the square-wave
quadrature generator allows optimum trimpot tuning, the procedure is still
not for the faint-of-heart: it should be attempted by experienced homebrewers.
Excellent sideband suppression
of 50 dB has been achieved with a little extra tuning. The resulting receiver
has contest-quality characteristics, with good dynamic range and few spurious
responses. The sharp cutoff audio filter can be set very wide (when listening
after a CQ) or narrow, for QRM elimination. A bandpass response would be
better, but would require another front panel control to set the centre
frequency. Break-in muting is excellent, when the time-constant is matched
with the transmitter.
