ALC

Today, the typical transceiver output is 100W to 200W. There are amplifier tubes that can be destroyed by 100W of drive. A good example is the 3CX800A7. Driving a 3CX800A7 with 100W PEP will eventually strip flakes off of the cathode. The flakes lodge between the cathode and the grid cage--creating a fatal short. Even a pair of 3CX800A7s are clearly over-driven by 100W. The fix is: connect a approx. 40 ohm cathode negative-feedback resistor in series with each 3CX800A7 cathode. As a result, the 3CX800A7s won't be driven above their maximum ratings--and into non-linearity--by a 100W transceiver. Naturally, when cathode negative-feedback resistors are added, the cathode driving impedance increases. The driving impedance for a pair of 3CX800A7s is about 25 ohms. With 40 ohm cathode resistors, the driving impedance is roughly 50 ohms.

Cathode negative-feedback resistors are better than having a matched pair of 3CX800A7s. The cathode currents automatically equalize themselves---and unlike ALC circuits, cathode feedback resistors work instantaneously--eliminating ALC's generic flaw--leading edge splatter on SSB. Amplifier-to-transceiver ALC works properly only on constant signal level modes such as RTTY and FM.

The 3-500Z is rated at approx. 60 watts drive. When a single 3-500Z is driven by 100W, it "flat-tops" and produces distortion. A 25 ohm low-L cathode feedback resistor will make a 3-500Z linear with 100W of drive. The resistor is placed in series with the cathode RF coupling capacitor.

Capacitors and RF Current

Capacitors that carry RF current are subject to two types of internal heating. Like ripple filter capacitors, the ESR in the capacitor's conductors generates heat that is proportional to I^2 x R. Due to skin-effect, R goes up with frequency. Another source of heat is RF dielectric loss. Since dielectric loss usually varies with frequency, the current carrying ability of capacitors changes with frequency. Typically, transmitting capacitors are current rated at three widely-spaced frequencies. It's a good idea to check the manufacturer's current ratings before using a transmitting capacitor in a specific application. Just because a capacitor is a transmitting-type does not mean that it will work reliably in all RF applications.

Tuned Circuit Q

Most of the tuned input and tuned output circuits in HF amplifiers are pi-networks. There are a number of ways to define the Q of a pi-network. In what follows, Q is defined as the input impedance of the pi-network divided by the reactance of the input, shunt element--typically a capacitor. This definition of Q is the one used by Eimac® in Care and Feeding of Power Grid Tubes.

Tuned Input Circuits for Class AB2 Cathode-Driven [grounded-grid]

Even though grounded-grid amplifier circuits look simple, they are not. The grounded-grid amplifier's tuned input circuit is in series with and out of phase with the anode current pulses. The RF cathode current's approx. half sine wave pulses are the sum of the anode and grid currents. Since the driver is connected to the other end of the tuned input, some of the RF cathode current finds its way back to the driver. Consequently the driver interacts with the amplifier. The Q of the amplifier's tuned input affects this interaction.

Modern solid-state output MF/HF transceivers use a broadband push-pull RF output stage. In order to meet FCC requirements, Butterworth and/or Chebyshev pass band filters are used to suppress spurious emissions. Such filters introduce inductive reactance or capacitive reactance within their pass bands. In other words, the output impedance of a modern transceiver is seldom 50 ±j0 ohms. When driving a tuned input in a grounded-grid amplifier, filter reactance interacts with the input reactance in the tuned input. The length of the coax between the driver and the tuned input affects the interaction.

When tube manufacturers state the cathode driving impedance in grounded-grid operation, they are talking about an average value. The instantaneous driving impedance fluctuates wildly during the sine wave input signal. During most of the positive half of the input cycle, the grounded-grid looks negative with respect to the cathode--so the flow of current is cut-off. Since virtually no current flows, the driving impedance is extremely high.

During the negative swing in the input cycle, the grounded-grid is relatively positive. A positive grid accelerates electrons away from the cathode, producing high anode-current and grid-current. Due to the large flow of current, the input-impedance is low during the negative half of the input cycle.

Consider a pair of 3-500Zs. When the driving voltage is peaking at negative 117v, the anode-current is at its peak, and the instantaneous anode-voltage is at its lowest point--about +250v. At this instant, the total, peak cathode-current is 3.4a. Thus, the instantaneous cathode driving impedance is 117v/3.4a = 34.5 ohm--and the peak driving power = 117v x 3.4a = 397W.

In other words, the instantaneous driving impedance swing is from near-infinite all the way down to 34.5 ohms. The instantaneous drive power requirement varies from 0w at the positive peak to 397w at the negative peak of the input sine wave. Thus, the input pi-network's job is to act as a flywheel/energy storage system and a matching transformer. That's why a simple broadband transformer can not adequately do the job of matching the driver impedance to the cathode impedance in a grounded-grid amplifier.

The Q of a tuned circuit is like the mass of a flywheel. More Q makes for a better flywheel--which does a better job of averaging the wild swings in input-Z--thereby producing a lower input-SWR. The trade-off is that more Q means less bandwidth. With a high Q, the input SWR may be near-perfect at the center of the band, but unacceptable at the band edges. Thus, a compromise is in order. Eimac® typically recommends using a pi input network Q of 2 for Class AB2 grounded-grid operation. To arrive at a Q of 2, the reactance [X] of the input capacitor, C1, is minus j50 ohm÷2=minus j25 ohm. Using C=1÷[25(2f)], approximately 220pF of input capacitance is needed for a Q of 2 on the 10m band. In actual practice, however, 220pF may be far from the value that produces a satisfactory SWR with a particular model transceiver and a particular length of coax. It may be possible to find a length of coax that would ameliorate this problem on 10m--but there are eight other bands to contend with below 30MHz. Since band switching different lengths of coax is hardly practicable, it would be useful if the input capacitors were adjustable in a grounded-grid amplifier's tuned input circuits. Adjustable coils are also useful.

Tank Circuits

When the Q of the output pi-network tank circuit is low, two problems can occur. The harmonic attenuation may be inadequate to meet FCC requirements--and the load impedance matching range decreases. In other words, when Q is low, the tank circuit may be incapable of matching even a 50 ohm load. When the Q of the tank is too high, efficiency decreases due to the increase in I^2 R circulating current losses. A compromise is in order. A Q of 10 is about minimum. A Q of 20 may cause excessive tank component heating due to high circulating current. A Q of 12 to 15 is a fair compromise.

Better tank performance can be achieved by using a pi-L tank circuit. When compared to a simple pi, the pi-L has roughly 15db better harmonic attenuation and it typically has a wider matching range. The trade-offs are that the pi-L requires an extra switch section and a tapped inductor.

Skin Effect and Current Capability

As frequency increases, progressively less current flows inside a wire--so current progressively concentrates on the surface. Since a steadily decreasing part of the conductor is being used, resistance increases as frequency increases. For example, a 12 gauge (copper) wire will carry 20A at 60Hz with very little heating. At 30MHz, the RF current carrying ability of 12 gauge wire is about 5A. Band switch contact current ratings need to be similarly de-rated as frequency increases. Paralleling contacts is a good way of increasing the current handling ability of a band switch. Directing a portion of an amplifier's cooling air flow at the band switch improves the RF current handling ability of band switch contacts.

HF tank inductors can become quite lossy unless the conductor surface area varies in proportion to frequency. Inadequate tank conductor size is the main reason for decreasing amplifier efficiency at the higher frequencies. A tank inductor made from 14 gauge wire is usually more than adequate for efficient 1.8MHz operation at 1500W PEP. For efficient operation at 29MHz, approx.10 mm o.d. copper tubing (or copper strap with an equivalent surface area) is appropriate. However, due to normal QSB--at the receiving end, even a one-third decrease in transmit power is virtually undetectable. Thus, squeezing out the last percentage of efficiency on 10m is not very important.

Calculating the RF circulating current in a tank inductor is fairly complex. A quick approximation is to multiply the maximum anode-current by Q. For example, if the anode-current is 1.2A and the tank Q is 15, the RF circulating current in the tank will be 1.2*15 =18A. At 29MHz, 18A is a formidable amount of current.

Silver

Compared to copper, silver [Ag] is cosmetically more attractive and more immune to oxidation. However, silver does not make an amplifier measurably more efficient at frequencies below about 100MHz. Copper oxidation can be prevented by polishing copper with extra fine steel wool and applying clear, gloss, polyurethane varnish.

Silver is useful as a component of solder. 95% tin [Sn], 5% silver, solder has a melting temperature of 221 degrees-C/430 degrees-F. Compared to tin-lead electronics solder, 95/5 Sn/Ag solder is about 3.5 times stronger and it has better wetability--especially on hard-to-solder materials. 95/5 Sn/Ag solder is ideal for soldering tank components, band switches, surface-mount solid state devices, loose vacuum tube pins, and low Q parasitic suppressors. When resoldering a tin-lead solder joint with tin-silver solder, first remove as much of the tin-lead solder as possible.

Anode HV RF Chokes

The basic requirements are: 1. The choke must have ample reactance at the lowest operating frequency to limit the RF current through the choke to a reasonable amount. 2. The choke can not be self-resonant near an operating frequency. 3. The wire gauge used must be able to carry the DC anode current plus the RF current at the lowest operating frequency without excessive heating.

If the HV-RFC has a self-resonance on or near an operating frequency, potentials of many times the anode supply voltage can appear on the choke. When this occurs, a choke arc and fire is likely. Choke fires can destroy more than just the choke because the rising plume of ionized gasses from the choke fire often creates a conduction path to the ceiling of the RF output compartment. If an arc occurs, pervasive damage is likely if no glitch protection resistor was used in the HV positive circuit.

Choke Design

Materials:
There are two types of wire insulation materials that are suitable for use in HV-RFCs--silicone varnish and Teflon. Modern, high-temperature electric motor wire is insulated with a tough, silicone varnish that can handle high DC voltage and high RF voltage. At room temperature, a twisted pair of #20 silicone varnished wires can withstand more than 5000VDC or 1500W in a 50 ohm circuit at 29MHz. This type of wire is sold by the pound in electric motor rewinding shops. If you want to buy some, bring your own empty spools and winding device--such as a variable-speed electric drill, with a homemade adapter to hold the spool. Due to its toughness, silicone varnish insulation requires a special method of stripping. An open flame from a butane lighter causes the silicone varnish to decompose and combust. The remaining ash residue can be removed from the copper with steel wool.

Teflon insulated magnet wire is not common. Although ordinary Teflon insulated hookup wire may be used, the extra insulation thickness requires that a longer coil form be used. One potential trade-off with Teflon insulated wire is phosgene. When Teflon burns, deadly phosgene [COCl₂] gas is produced.

Due to contact with air, the current carrying ability of either type of wire is much higher in an HV-RFC than it would be in a transformer. #28 wire will easily carry 1A in a HV-RFC. #24 will carry several amperes with acceptable heating.

G10 or G11 epoxy-fiberglass tubing is RF-resistant, strong, and easy to work with. It is an ideal material for building HV RF chokes. It can be obtained from plastic supply houses. 1mm wall thickness is more than adequate. Diameters of 16 to 25 mm are typically used for building HV-RFCs. G10 tubing can be cemented to a G10 base plate with silicone rubber adhesive or epoxy. A source of G10 tubing: Plastifab,1425 Palomares, La Verne, CA 91750 818 967 9376.

It is probably a good idea to limit RF current in the HV-RFC to no more than 1 ampere. To calculate current in the choke, take roughtly 2/3 of the anode supply volts and divide it by the reactance in ohms at the lowest operating frequency -- a.k.a. Ohm's Law.

Determining Bypass C

Power supply components can be damaged by RF. Electrolytic filter capacitors are especially at risk. Thus, adequate RF bypassing on the power supply side of the HV-RFC is needed. Probably no more than 10V of RF should be allowed to appear on the +HV supply at the lowest operating frequency. Determining just how much bypass C is needed basically involves using ohm's Law. The amount of RF current flowing through the choke and the amount of bypass C need to be evaluated for the lowest operating frequency--usually 1.8MHz. For example, if the reactance of the choke is +j2000 ohms, and the AC anode voltage is 2000Vrms, then I=2000V/2000 ohm=1A of RF flows through the choke. In order to limit the RF voltage to 10V maximum at 1.8MHz, 10V/1A=10 ohm of capacitive reactance is needed for an adequate bypass. Using C=1/(Xc * 2pi * f), this equates to a HV bypass capacitance of 8842pF. Obviously, a typical 1000pF bypass C [minus j88 ohm] is not going to do the job because it would allow approx. 88V of RF to appear across the HV supply if 1A were flowing through the choke.

500pF 20kV TV-type doorknob capacitors are NOT designed to handle RF current--so they do not make satisfactory HV bypass capacitors. Disk ceramic capacitors may be used for HV bypassing. Disk ceramic capacitors are somewhat limited in the amount of RF current they can safely handle. Manufacturers typically don't publish RF current ratings for them. To find out how different capacitors react to RF current, you must test them yourself. Even a 7500WVDC, 2500pF disk ceramic capacitor becomes warm from 1A at 1.8MHz. Thus, it is often best to parallel a number of individual bypass capacitors so that the RF current will be shared among them.

Determining L

At the lowest operating frequency, the HV-RFC should have enough reactance to limit the RF circulating current through the choke to a reasonable amount. Allowing a RF current of 1A RMS through the choke usually does not create problems for the wire-lead disc-ceramic capacitors that are typically used to bypass RF on the power supply side of the HV-RFC. To minimize RF current through the choke, it would seem that more inductance is the answer. However, more inductance means more choke resonances and a greater likelihood of choke fires. A compromise is indicated.

Over the years, various schemes have been used to minimize choke resonances. Adding gaps at presumably esoteric positions in the winding was represented as a means of decoupling parts of the choke winding--allegedly ameliorating the self-resonance problem. However, when the resonances of gapped chokes are compared to similar chokes without gaps, no real improvement is observed on a dipmeter. This should not be surprising. Optimum decoupling between two coils occurs when they are mounted at a right angle. Adding end-to-end spacing with gaps is the least effective decoupling method possible. To minimize resonance problems, instead of using a single large choke, use two smaller chokes mounted at right angles.

The highest-L choke that can built that is free of self-resonances in the HF spectrum is roughly 60µH. At 1.8MHz, 60µH has a reactance of about +j679 ohm.

The RMS voltage that appears across an amplifier's HV-RFC is approximately two-thirds of the anode supply voltage. For example, an amplifier that is powered by a 3000V supply subjects its HV-RFC to about 2000V RMS. If a 60µH inductor was used in this amplifier, at 1.8MHz the RF current through the choke would be 2000V/679 ohm=2.95A RMS. Adequately bypassing approx. 3A of current on the power supply side of the choke is difficult. A typical HV disk ceramic bypass capacitor can handle only about 1A. Another problem is that at 1.8MHz 130pF [minus j679 ohm] of extra capacitance is required from the tune capacitor to cancel the +679 ohms of reactance in the choke. Adequately bypassing 3A at 1.8MHz requires a substantial amount of capacitance. To hold the voltage across the bypass capacitors to less than 10V at 1.8MHz, roughly 0.026µF [minus j3.3 ohm] is indicated. To handle this amount of current, four approx. 0.0075µF HV disc ceramic capacitors would probably be needed. All things considered, using more inductance is indicated. Limiting the HV-RFC's RF current to a maximum of !A would make the task of bypassing a lot easier. However, increasing the inductance above 60µH is virtually certain to move choke resonances into the HF range. Unless these resonances are prudently parked between operating frequencies, a choke fire may result.

To realistically evaluate the self-resonance situation, HV-RFCs should be checked with a dipmeter after they are installed and wired in the amplifier. If a self-resonance is within about 5% of an operating frequency, there may be a problem. When re-parking resonances, it is usually best to remove turns from the choke. This will move the resonances up in frequency--and only slightly increase the maximum RF current through the choke.

In continuous coverage amplifiers, there are obviously no safe parking places for choke resonances. The only solution is to switch HV-RFCs with one or more HV vacuum relays.

HV-RFCs should be single-layer solenoid wound. To minimize wire vibration during operation, the wire should be under constant tension when winding and soldering the ends to the solder lugs. When silicone varnish insulated wire is used to wind a HV-RFC, the finished winding should be given a coat of gloss urethane varnish to hold the wire in place. Since varnish will not adhere to Teflon wire, a different method is needed to keep a Teflon winding taught. Small tensioning springs are soldered to the ends of the wire. The springs provide constant pull to minimize wire vibration during modulation. An S-shaped copper foil jumper should be connected across each tensioning spring.

DC Blocking Capacitors

Blocking high voltage DC is the least difficult part of the blocking capacitor's job. During operation on 10m, the DC blocking capacitor must be able to carry most of the RF circulating current in the tank. Here's why: The amplifier tube's anode capacitance normally provides most of the tune capacitance during 10m operation. Thus, a major portion of the tank circulating current passes through the anode capacitance and therefore through the DC blocking capacitor. In an amateur radio amplifier, blocking capacitor currents of 5 to 10 A RMS are not uncommon during operation on the 10m band.

Selection of a blocking capacitor should not be guesswork. It is advisable to select a capacitor or capacitors that is rated to carry the calculated maximum RF current present. Merely selecting an RF-type (transmitting) capacitor is not good enough. Some RF-type capacitors have rather unspectacular current capabilities. The capacitance of the DC blocking capacitor is not very critical. 1000pF seems to be more than adequate for operation at 1.8MHz. 88 ohms of Xc is relatively insigificant in comparison to the typical 1000 to 2000 ohm anode output Z.

Vacuum Components

Vacuum capacitors and vacuum relays are ideal for use in high power RF amplifiers because they can withstand high RF voltages. vacuum capacitors are able to handle more RF current than any other type of capacitor. There are some trade-offs. Vacuum components depend on their glass-to-metal or ceramic-to-metal seals to maintain their near-perfect vacuum. If a seal leaks, air molecules enter and the vacuum component is kaput. Vacuum component seals should not be subjected to unnecessary mechanical stress.

Vacuum Capacitors

Although vacuum capacitors can be mounted in any position, vertical mounting places the least stress on the soft copper plates. Vertical mounting also makes the most efficient use of chassis space. With vertical mounting, a right-angle drive is used to bring the 1/4" diameter tuning shaft to the front panel. Cardwell-Multronics® makes a compact right-angle drive mechanism that is ideal for this application. It is designed to replace the shaft-cap on a vacuum capacitor's tuning shaft. The vacuum capacitor should be set for minimum C before the drive shaft cap's setscrews are loosened.

A vacuum capacitor should not be used as a standoff-insulator to support heavy components. High G force can be fatal to a vacuum capacitor. The danger is not necessarily breakage or damage to the seals. The plates in a vacuum capacitor consist of a series of concentric, intermeshing, soft copper cylinders that almost touch each other. A vacuum capacitor can be shorted by an inertia force that is capable of bending the soft copper plates.

Vacuum Relays

To avoid stressing the seals, connections to the contact terminals of vacuum relays should be made with soft copper ribbon.

The molded-in coil terminals on vacuum relays are easily broken. Connections to the coil terminals should be made with approx. 24 gauge stranded hookup wire.

Vacuum relays generate sharp mechanical vibrations when they switch. If one is mounted securely to the chassis, the chassis acts like a speaker cone--coupling the vibrations more efficiently to the air. One way of overcoming this problem is to mount the vacuum relay on small beads of silicone rubber. To accomplish this, drill a approx. 3mm oversize mounting hole in the chassis. Use temporary L-shaped poster board spacers to prevent the relay from touching the chassis. After cleaning the surfaces with acetone, apply three small beads of silicone rubber between the relay mounting flange and the chassis. Allow the silicone rubber to cure for 2 days. Remove the spacers. The relay should float quietly on silicone rubber shock absorbers. The vacuum relay's body should be grounded to the chassis with thin copper ribbon. The ribbon may be soldered to the edge of the relay flange. To avoid overheating the seals, use a large soldering iron--and tarry ye not.

Controlling Make and Break Times

All relay coils have inductance. Since inductance delays a change in the flow of current, coil-inductance tends to increase the make-time of relays. Make-time is an important design consideration when using vacuum relays for RF switching. RF-rated vacuum relays use copper contacts to obtain high conductivity. However, copper is vulnerable to damage from hot-switching. For example, if an amplifier's RF output relay contacts are not closed and finished bouncing before the RF arrives, arcing and contact damage is likely.

Make-time can be decreased by supplying extra voltage to the coil during start-up with what is commonly called a speed-up circuit. Jennings® and Kilovac® recommend using them to accelerate relay closure. A speed-up circuit consists of a resistor in series with the relay's coil and a power supply that supplies two to three times the rated coil voltage. At turn-on, the extra voltage hastens the flow of current in the coil. The resistor limits the steady-state coil voltage to a safe value after the flow of current builds up in the coil.

DC relay coils are usually paralleled with a diode to absorb the reverse voltage spike that results when current stops flowing through the coil. If no reverse diode is used, the reverse voltage spike can exceed 20 times the rated coil voltage. The break-time of a DC relay can be controlled by adding a resistor in series with the diode. As R increases, the break-time decreases. R should probably not exceed three times the coil resistance.

Testing Vacuum Components

When a vacuum seal leaks air, the breakdown voltage decreases. This problem is easy to spot in a glass-body vacuum relay--because when electrons flow through air, blue-purple photons are emitted. With glass-body vacuum capacitors, this problem is not as obvious. In a leaky glass-body vacuum capacitor, internal ionization/arcing is often not visible since the problem usually occurs deep inside the meshed concentric plates.

It is a good idea to test all vacuum components, whether they be new or used,before constructing the amplifier.

When a vacuum component in an amplifier becomes gassy, arcing typically occurs near the crest of the RF sine wave--so a bad vacuum component typically reduces the peak power output. Since many amplifiers use more than one vacuum component, finding the bad one is difficult without individual evaluation using a breakdown tester.

Random replacement--a.k.a. "Easter-egging"--is not an efficient way to repair an amplifier that uses vacuum components. For instance--if an amplifier's RF output vacuum relay becomes gassy, it is virtually certain to divert high power RF into the (usually more delicate) RF input relay. If a thusly-damaged RF input relay is replaced, the new RF input relay may also be damaged by the gassy RF output relay. Thus, it is desirable to be able to individually test vacuum components with a breakdown tester.

Testing the quality of a vacuum is similar to testing the breakdown voltage of a diode. Connect a approx. 100M ohm HV resistor and a approx. 20 microampere meter in series with a breakdown tester. Increase the voltage until about 1 to 2 µA of leakage is detected. This voltage is the breakdown voltage. The peak RF working voltage of a vacuum component is roughly 60% of the DC breakdown voltage.

Measuring Relay Contact Resistance

There are basically two types of vacuum relays--those that are designed for hot-switching, and those that are not. Hot-switching-capable relays have tungsten contacts. Such relays are intended for use primarily in power supplies. Relays that are designed for RF have copper contacts. They should never be allowed to hot-switch. Copper-contact relays have approximately one-third the contact resistance that similar tungsten-contact relays have. For instance, the Jennings RJ-1A is the copper-contact version of the tungsten-contact RJ-1H. The rated contact resistance of the RJ-1H is 30 milli-ohms. The rated contact resistance of the RJ-1A is 10 milli-ohms. Tungsten contact relays are not rated for current RF current. However, they should work fine for RF service if they are operated at roughly two-thirds of the RF current rating for their copper-contact counterparts. Tungsten contacts are extremely hard. They are capable of more operations than copper contacts. For heavy , full break-in telegraphy use, tungsten contacts are preferable--even though they do not have the continuous RF current handling capability of copper contacts.

With vacuum relays, contact failure is not uncommon. Contacts suffer from contact erosion. This condition increases contact resistance. Eventually, an eroding contact will open completely. To test a vacuum relay, the resistance of normally open [NO] contacts and the resistance of normally closed [NC] contacts should be measured and compared with the manufacturer's specifications. Ordinary ohm-meters are not suitable for detecting contact problems other than an open circuit. The voltage drop across relay contacts should be measured with a substantial current flowing. 1A is a reasonable current to use. Measure the mV drop directly across the contact terminals using a DMM with test prod leads. Most of the vacuum relays that are designed to handle RF current have a rated contact resistance of less than15 milli ohm--so no more than 15 milli V should appear across the terminals with 1A flowing through the contacts.

Testing Vacuum Capacitors

Vacuum capacitors store energy efficiently because they have virtually zero ESR [equivalent series resistance] and internal L--thus, the peak discharge current can be astronomical. When the breakdown test voltage is high enough to create more than a few microamperes of leakage, a vacuum capacitor will normally self-discharge--producing a clearly audible tick due to the large peak discharge current and commensurately large electromagnetic force. After a vacuum capacitor self-discharges, it begins charging and the process repeats. A vacuum capacitor should not be allowed to self-discharge more than a few times unless the capacitor has been in storage for many years. During long-term storage, for some as yet unexplained reason, copper atoms tend to line up, forming whiskers on the surface of the plates. Copper whiskers initially reduce the breakdown voltage. Copper whiskers can be dislodged by self-discharge. If the breakdown voltage increases after a self-discharge, another self-discharge may be beneficial. Repeated self-discharge will cause a decrease in breakdown voltage.

Grounded-Grid Amplifier Tune-up

Linear amplifiers are like induction motors--they are designed to run fully-loaded. If your grounded-grid amplifier's instruction book says to reduce drive power during tune-up--and most of them do--it is not giving you correct information. In order to be linear, amplifiers must be tuned-up with the same peak drive-power level that they will be driven with during actual operation. Reducing drive power changes the output Z of the amplifying device to something other than the tank circuit was designed to match. Thus, the tune and load settings with low drive will be wrong when normal drive is applied.

Tune-up method #1: Set the amplifier's HV supply to the CW-Tune/low-V-tap. If you are not sure where to preset the Load control, set it to >70% of maximum loading [30% of C] to be safe. Apply the drive level that you intend to drive the amplifier with during actual use. Alternately adjust the amplifier's Tune and Load controls for maximum relative power output. The whole process should take less than 6 seconds. It may sound brutal, but this tune-up method results in good amplifier linearity and it won't damage the tubes if the maximum anode-current rating is not exceeded. If the anode-current is excessive, the resistance of the cathode, RF negative-feedback resistor needs to be increased slightly--or the PEP adjust control in the transceiver needs to be turned down.

Tune-up method #2: [not for FM, AØ, and RTTY operation] To reduce the stress on an amplifier during tune-up, use a reduced duty-cycle driving signal. This can be accomplished by keying the transceiver, on CW mode, with a CW keyer, set to send approx. 50wpm dits. CW dits have a 1/2-on, 1/2-off, or 50% duty-cycle. Using this method, the amplifier may be tuned-up, again for maximum power output, in its higher-voltage, SSB-mode. Keyers that have a weighting adjustment can be set to produce a light dit that has a duty cycle of about 30% instead of the normal 50%. Another device for reducing the duty-cycle during tune-up is a tuning-pulser.

If you want to operate with reduced power during good band conditions, first tune up your amplifier with normal drive power, then turn the microphone gain down to reduce power.

Class AB1 Design Considerations

Choosing the Optimum Value of Grid Terninating Resistance.

Tetrodes and pentodes require a peak RF drive voltage that semi-matches up to the grid bias voltage.

My strategy is to choose a value of grid termination resistance that roughly provides the needed peak RF drive V to the grid with exciters that develop 100v-p (100w rms) to 141v-p (200w rms) across 50 ohms. In other words, the goal is to match peak RF drive volts with the needed grid bias volts from the tetrode/pentode manufacturer's technical specifications.

• -- For tubes that require 50v to 70v of grid bias, like the 4CX800A and 4CX1000A, a voltage-halving bifilar stepdown transformer driven 12.5 ohm grid termination is used.
• -- For tubes that require 100v to 140v of grid bias, like the 4CX1500A, a directly-driven 50 ohm grid termination would be used.
• -- For tubes that require 200v to 280v of grid bias, like the 4-1000A and 8169, a voltage-doubling bifilar stepup-transformer driven, 200 ohm grid termination is used.
• -- For tubes that require greater than 300v grid bias, like the 8171, a trifilar stepup-transformer driven, 450 ohm grid termination is used.

However, if the peak grid V with max. drive is still a bit much, a cathode RF negative feedback reisistor (Rk) can be added to make up the difference. However, a trade-off is that the peak V drop across Rk normally subtracts from the screen to cathode V at the critical anode current peak. A workaround is to use the circuit shown in Figure 10.

Tuned Input Circuits for Neutralized, Class AB1 Grid-Driven Operation

Class AB1 grid-driven amplifiers look more complex than Class AB2 grounded-grid amplifiers. However, the tuned input circuitry for multi band Class AB1 grid-driven operation is comparatively simple.

The grid capacitance of tubes that are commonly used in Class AB1 grid-driven amateur radio power amplifier service ranges from about 15pF to 130pF. Since the capacitance of the grid is in parallel with the input, as frequency increases, input SWR worsens. This problem can be corrected by connecting a variable inductor in parallel with the grid. The inductive reactance {+j ohms} of the inductor is adjusted to cancel the capacitive reactance {minus j ohms} of the grid--thereby resonating the grid at the operating frequency. When the input SWR is tuned to minimum, the grid circuit is resonant. A simplified diagram is provided.

If the other end of the variable inductor is connected to a properly-adjusted capacitive voltage divider (connected between the anode and chassis ground), the amplifier is neutralized at whatever frequency the grid is tuned to. Obviously, this type of Class AB1 input circuit is a natural for continuous HF and MF coverage--just what's needed for operation on the 9 amateur bands below 30MHz. The ratio of the capacitances in the capacitive voltage divider equals the ratio of the feedback capacitance (the anode to grid capacitance) divided by the grid input capacitance. Typical ratios are 150 to 1 ... Achieving wide frequency coverage is not as easy in Class AB2 grounded-grid operation. A pi-network tuned input with the recommended Q of 2 has a limited bandwidth--so many, switched, tuned input circuits are required for wide frequency coverage.

Screen and Grid Supplies

There are many tetrodes and pentodes to choose from that are satisfactory for Class AB1 grid-driven operation. The essential criteria is that, with zero grid volts, the tube is capable of a peak anode-current that is at least triple its maximum (average) current rating. In most cases, this condition can only be met if near-maximum screen-voltage is applied. Relatively high screen-voltage is important because peak anode-current is a function of the screen-voltage raised to the 1.5 power.

For the best linearity, screen voltage should be regulated. For smaller tetrodes and pentodes, a Zener diode shunt regulator offers a good solution. Typically, a series of 10v to 30v, 5W Zeners are used. Screen voltage is adjusted by shorting out Zener diodes with a rotary switch. For larger tubes, an adjustable series-regulator is the best way to supply voltage to the screen. Thanks to modern power FETs and the venerable 723 IC linear regulator, building a reliable, regulated supply of 2kV or less is fairly simple.

Since the grid does not pass current in Class AB1 operation, there is no necessity to regulate the bias voltage. However, the bias supply should not have an extremely high output impedance. A maximum grid circuit R of 1k to 100k ohms is typically recommended by tube manufacturers.

Work Space

'Work-space' and 'head room' are terms that describe the range in which instantaneous anode-voltage is free to move up and down--thereby performing work. In a tetrode, at the maximum peak anode-current, to avoid excessive screen-current and a decrease in linearity, the instantaneous anode-voltage should not dip much below the screen-voltage. For example, a tetrode with a 4kV anode supply and an 700V screen supply, the work-space is approximately 4000V minus 600V = 3400V peak

In a pentode, the instantaneous anode-voltage may dip close to the suppressor-voltage--which is typically zero volts. In the above example with a screen-voltage of 800V, if the tube happened to be a pentode, the work-space would be around 3750V peak. Thus, pentodes enjoy slightly more work-space than tetrodes. As a result, pentodes are slightly more efficient than tetrodes. However, pentodes are more expensive than tetrodes because they are more complex to build. Sockets with low-L suppressor and screen bypass capacitors are needed for stable operation. Pentode sockets are not inexpensive. Another trade-off is that there are relatively few types of pentodes to choose from. A (if not the) suitable pentode for amateur radio Class AB1 grid-driven service is the 5CX1500.

Pentode Caveats

Pentodes typically have less feedback capacitance than tetrodes. This advantage theoretically makes pentodes more stable. Some designers do not neutralize pentodes because they feel the relatively low feedback capacitance between the anode and the grid is insignificant. However, for optimum linearity and stability, plus low input SWR, a pentode should be neutralized. This can easily be accomplished with the grid input circuit diagram [Figure 5] for Class AB1 tetrodes. To use this circuit with a pentode, DC-connect the suppressor to the cathode with a l0 or so ohm resistor. However, the suppressor must always be RF-bypassed to chassis ground to decrease feedback from anode to grid.

Screen Protection

Every screen type tube has a maximum screen dissipation rating in watts. If screen current times screen voltage exceeds this rating, the tube could be destroyed. This can easily happen with a no load or light load condition--so various protection schemes are used. If the anode voltage disappears while screen voltage is present, screen current will be excessive unless a means of protection is provided. Another hazard is reverse screen current. Reverse screen current can easily become a runaway condition. It happens virtually instantaneously. Reverse screen current is commonly experienced in Class AB1 operation. Unless bled off into a resistor load or into a shunt Zener voltage-regulator, reverse screen current can quickly destroy a tube. For tubes with screen voltages in the 300V to 800V range, a shunt regulator using a Zener diode string is a good solution. The Zener regulator string is connected through a high-R resistor to the anode supply. A sample circuit is provided. A suitable tube would be the 4CX1500B, or similar types.

Advantages of Shunt Zener Screen Regulation:

• Limits the maximum current that can be drawn by the screen.
• Protects against reverse screen current.
• If the high voltage disappears, so does the screen voltage.

However, for larger tubes with higher screen current and screen voltage requirements, a Zener shunt regulator is somewhat impractical. A continuously-adjustable series-regulator screen supply is a better solution. To protect against reverse screen current, a shunt resistor/bleeder must be connected across the screen supply. A bleeder current flow of roughly 20% of the normal screen current seems to be adequate. 25% might be safer. To protect against excessive forward screen current, a fast acting fuse or magnetic-type circuit breaker is incorporated in the primary of the screen supply power transformer. An adjustable series regulator circuit is provided.

Grid-Driven Class AB1 Amplifier Neutralization and Tuneup

ßAdjusting a Class AB1 amplifier may look complicated at first, but after you have done it a few times, and you begin to understand the reason behind each step, it gets easier.

Neutralization: The goal of neutralization is to isolate the anode from the grid at the operating frequency. Neutralization discourages regeneration--oscillation. Neutralization usually needs to be adjusted only once.

1. Disconnect the amplifier from the electric-mains.

2. Temporarily disconnect the tank circuit from the HV blocking-capacitor.

3. Substitute a low-L film resistor, with the same R as the design anode-load [output] resistance, in place of the tank circuit. Typical values would be 1000 ohm to 4000 ohm, 2W. The resistor connects to the blocking-capacitor and to chassis-ground. Connect an RF-voltmeter or an oscilloscope equipped with a 10 to 1 hign impedance probe across the resistor.

4. Connect the amplifier to the electric-mains and turn on the transmit-receive relay power supply plus the grid and filament supplies. Do not turn on the screen or anode supplies.

5. Drive the amplifier with 20m or 15m RF. Tune the grid-circuit variable-inductor [L1] for minimum input SWR or minimum reflected power. If necessary, adjust the DC grid-voltage so that virtually no grid-current flows.

6. Adjust the neutralizing-capacitor (C3) for minimum RF-voltage at the anode-load resistor. If needed, readjust L1 for best input SWR followed by readjustment of C3. This completes the neutralizing procedure.

After C3 is nulled, the amplifier is neutralized for all bands. To confirm this, check the neutralization on another band. Readjust L1 for minimum SWR. The RF voltage across the output load resistor should not change appreciably. Typically, no further adjustment is necessary--even if the tube is replaced.

Remove the resistor and reconnect the tank circuit.

Tune-up.
1. Switch off the screen and HV anode supplies. Switch on the T/R relay supply, the filament supply and the grid supply.

2. Transmit on CW-mode into the amplifier and adjust L1, the grid roller-inductor, for minimum input reflected power. This tunes out the grid-reactance and simultaneously neutralizes the amplifier at the operating frequency. If you are using a transistor-output transceiver, to preclude SWR shutdown, initially tune the grid with no more than 5W of signal.

3. Apply full drive power using an electronic keyer sending dits at about 50wpm, or a use a tuning-pulser. Adjust the DC grid-voltage so that <0.1mA of grid-current flows. The grid-voltage is adjusted so that the grid is on the threshold of current flow. The grid-voltage adjustment is not used to set the zero-signal anode-current [ZSAC]--also known as 'idling current' or 'resting current'. Although the grid-voltage adjustment can discretely be used to make a small adjustment in the ZSAC, in Class AB1 operation, the primary criteria for setting the grid-voltage is that virtually NO grid-current flow with maximum drive. ZSAC is set by adjusting the screen voltage. Switch on the screen and HV supplies. Key the amplifier but do not apply drive power. Using the screen voltage adjustment, set the ZSAC as recommended by the tube manufacturer. For most tubes, the ZSAC should be about 20% of the rated anode current.

4. If a variable tank inductor and a variable tune capacitor is used, preset the tune capacitor and the tank inductor for the desired operating Q on the band in use. Preset the load capacitor and inductor by calculation. It is best to error on the side of too-little load C [heavy loading]. If too-light loading [too much load C] is used, excessive screen-current is likely. Remember that the tune C sets the operating Q. Most of the tuning should be done with the variable tank inductor. Fine tuning can be done with the tune C--but the final setting should no be very far from the setting for the correct operating Q.

5. When any amplifier is tuned-up, the anode-current must be driven to the maximum, peak, design value so that the tube's output load resistance will meet the design criteria for the pi output tank circuit. If a lesser current is used without proportionately decreasing the supply voltage, the output load resistance will be too-high and the subsequent adjustment of the tank will be for the incorrect output load resistance.

Distortion

Perfectly linear amplification produces nothing except a larger representation of the input signal. Non-linear amplification produces mixing--and mixing creates distortion products.

Inter-modulation distortion [IMD] is the result of mixing between two or more input signals. The human voice produces many frequencies at any instant. When voice modulation is amplified non-linearly, many mixing products are produced. This is called "splatter" or, more descriptively, "rotten splatter." IMD is usually measured by simultaneously applying two equal-amplitude, not harmonically related modulation frequencies such as 2000Hz and 2200Hz. When two or more frequencies mix they produce spurious signals at their sum and their difference frequencies--in this case 4200Hz and 200Hz. The first level of mixing produces what are called "third order products." Additional products are produced by third order products mixing with the two fundamental frequencies. For instance, 2200Hz and 4200Hz mix to produce a signal at 6400Hz.

When distortion products are inside the fundamental pass band of an AM or SSB transmitter, audible distortion results. This gives voice modulation a rough, unpleasant characteristic that reduces intelligibility. Odd-order distortion products which lie outside the pass band can cause interference on adjacent frequencies.

There are two methods of referencing IMD measurements. In method A, the IMD power level is referenced to either one of two equal-amplitude input signals. The power ratio of PEP to either of two equal-amplitude sine waves is four to one [6db]. In method B, the IMD level is referenced to the PEP level. Thus, an IMD level of minus 34db using method A equals an IMD level of minus 40db using method B. Amateur radio operators tend to use method B because receiver S-meters respond to PEP. In commercial radio, the military, and the FCC--where distortion measurements are typically made with a spectrum analyzer--method A is used. When using a spectrum analyzer, distortion can be broken down further into third order products, fifth order products, seventh order products. However, total IMD referenced to PEP is a more significant number.

It is possible to measure IMD without expensive laboratory equipment. All that's needed is a receiver and some understanding of what's required to make a meaningful measurement.

By comparing the signal strength in the transmitter's fundamental pass band window with the signal strength in the adjacent pass band windows, IMD can be measured fairly accurately--even over the air. The amount of receive frequency offset is critical. If the receive pass band is too close to the transmitter's fundamental pass band, the receiver will not be able to separate the IMD energy from the fundamental energy. As a result of this overlap, the distortion measurement will be higher than the actual amount. If the receive frequency offset is too far from the fundamental pass band, the receiver's pass band will not receive all of the IMD--and the distortion measurement will be lower than the actual amount.

For a receiver with two, cascaded SSB filters, a receive offset of 3.6kHz is about right--provided that the receiver is set to the same sideband as the transmitter. For a receiver with one SSB filter, an offset of about 4.5kHz is needed. To measure the IMD level of a LSB signal, offset LSB-receive higher in frequency. For measuring the IMD from an USB signal, offset USB-receive lower in frequency.

Since very few S-meters are linear, a calibration chart of S-meter readings versus decibels is a prerequisite for making accurate measurements. A calibration chart can be made with a step-attenuator and a signal source, or with a signal generator/attenuator.

In order to measure IMD, at least two modulation frequencies are required. Human speech is a good signal source for measuring IMD because, at any instant, speech contains many fundamental frequencies and harmonics. As its name suggests, another harmonic-rich signal source is a harmonica. By simultaneously blowing into two or three adjacent holes at the low note end, a plethora of frequencies can be produced that are optimal for making distortion measurements.

Splatter Reports

Before reporting splatter, it is important to keep in mind that all SSB, DSB, and AM signals have IMD. In other words, everybody splatters. The obvious question is how many decibels down is the IMD? Minus 40db is excellent; minus 30db is objectionable; minus 20db is abundantly abominable. With one exception, FCC rules allow virtually any level of IMD inside the ham bands. The exception is when IMD causes harmful interference to emergency communications. Splattering on non-emergency communications is NOT considered to be harmful.

Before reporting a station's level of IMD, it is advisable to determine whether or not the station operator is interested in hearing your report. Although most amateur radio operators are interested in transmitting a high quality signal, some operators deliberately misadjust their equipment to maximize IMD.

Notes on Measuring Power

Since E-peak = E-rms x 2^0.5, and P = E^2 ÷ R, at its crest, the instantaneous peak power in a sine wave is double the RMS power. A common unit of measuring amplifier output is the PEP [peak envelope power] watt. Despite the name, peak envelope power watts are not peak watts--they are RMS watts at the crest of modulation. If an amplifier was powered by a regulated anode supply, there would be virtually no difference between PEP watts and AØ [NØN] watts. In a typical amplifier, the anode-voltage sags appreciably under AØ conditions--so PEP watts are typically about 20% higher than AØ watts. PEP need not be measured with voice modulation. PEP can also be measured by keying the driver at 30 pps with a steady string of pulses that approximates the duty-cycle of a human voice--roughly 30%.

Tube Ratings

Traditionally, amateur radio operators have taken a cavalier attitude toward tube manufacturer's ratings. While some ratings can be exceeded judiciously, exceeding other ratings can be costly. Examples of ratings which should not be exceeded for indirectly-heated cathode tubes are minimum filament-voltage and maximum anode-current. Violation of either rating can result in destruction of the delicate cathode. Directly-heated cathodes are more rugged. The maximum anode-current rating for directly-heated cathode tubes is a linearity issue--not a cathode destruction issue. One rating which should not be exceeded is maximum seal temperature. It has been said that the way to tell when the blower is too big is if it blows the tube out the socket.

END OF PART 4